High Efficiency Power Amplifier Architectures for RF Applications

ABSTRACT

A parallel delta sigma modulator architecture is disclosed. The parallel delta sigma modulator architecture includes a signal demultiplexer configured to receive an input signal and to demultiplex the input signal to output a plurality of streams, a plurality of delta sigma modulators executing in parallel, each delta sigma modulator configured to receive a stream from the plurality of streams and to generate a delta sigma modulated output, and a signal multiplexer configured to receive a plurality of delta sigma modulated outputs from the plurality of delta sigma modulators and to multiplex together the plurality of delta sigma modulated outputs into a pulse train.

CROSS REFERENCED APPLICATIONS

This application claims priority under 35 U.S.C. 119(e) to U.S.Provisional Patent Application Ser. No. 62/425,035, entitled“Switch-mode Power Amplifier (PA) Architecture for RF Application”,filed Nov. 21, 2016 and U.S. Provisional Patent Application Ser. No.62/427,641, entitled “Switch-mode Power Amplifier (PA) Architecture forRF Application” filed Nov. 29, 2016, and that are hereby incorporated byreference in their entirety as if set forth herewith.

FIELD OF THE INVENTION

The present invention relates to delta sigma modulators and morespecifically to parallel delta sigma modulator architectures forimproved power conversion efficiency in power amplifiers.

BACKGROUND

A radio frequency power amplifier (RF power amplifier) is a type ofelectronic amplifier that converts a low-power radio-frequency signalinto a higher power signal. There are many classes of power amplifierswhich are used to distinguish the electrical characteristics and methodsof operation of the power amplifiers. Accordingly, the classes of poweramplifiers are mainly lumped into two basic groups. The first are theclassically controlled conduction angle amplifiers forming the morecommon amplifier classes of A, B, AB and C, which are defined by thelength of their conduction state over some portion of the outputwaveform, such that the output stage transistor operation lies somewherebetween being “fully-ON” and “fully-OFF”.

The second set of amplifiers are the newer so-called “switching”amplifier classes of D, E, F, G, S, T among others, which use digitalcircuits and pulse width modulation (PWM) to constantly switch thesignal between “fully-ON” and “fully-OFF” driving the output hard intothe transistors saturation and cut-off regions.

Different types of power amplifier architectures may include differenttypes of components. For example, the type S power amplifiers convertanalogue input signals into digital square wave pulses by a delta sigmamodulator, and amplifies them to increases the output power beforefinally being filtered by a band pass filter.

In particular, delta sigma modulation is a method for encoding analogsignals into digital signals as found in an analog to digital (ADC)converter. Delta sigma modulation may also be used to transfer highbit-count low frequency digital signals into lower bit-count higherfrequency digital signals as found in digital to analog (DAC) operation.This technique is popular in modern electronic components such asconverters, frequency synthesizers, switched-mode power supplies andmotor controllers, due to its cost efficiency and reduced circuitcomplexity.

In addition, delta sigma modulators may reduce noise using noise shapingand increase signal resolution using filtering. In noise shaping, noiseis filtered by a noise shaping filter. This means that the noise isreduced inside frequencies of interest and increased outside thefrequencies of interest. As a result, the resolution of the signal isincreased. In delta sigma modulators, noise shape filtering may beperformed at an over-sampled rate. The noise shaping is achieved bysubtracting estimated in-band noise from an input signal of the deltasigma modulator. The estimated in-band noise subtraction is done throughthe feed-back path in the modulator. A post noise shaping filter can beplaced after the modulator that cuts the noise from outside thefrequency of interest which, in turn increases the signal's resolution.

As such, delta sigma modulators can provide a less complex and costefficient manner to perform Analog to Digital (A/D) and Digital toAnalog (D/A) conversion in many electronic components including, but notlimited to ADCs, DACs, frequency synthesizers, switch-mode powersupplies, and motor controllers.

SUMMARY OF THE INVENTION

Systems and methods in accordance with embodiments of the invention useparallel delta sigma modulators for improved power conversion efficiencyin power amplifiers. In accordance with one embodiment, a parallel deltasigma modulator includes a signal demultiplexer configured to receive aninput signal and to demultiplex the input signal into several streams ofsymbols at symbol boundaries; several delta sigma modulators, where eachdelta sigma modulator is configured to receive a stream of symbols fromthe several streams of symbols and to generate a delta sigma modulatedoutput; and a signal multiplexer configured to receive several deltasigma modulated outputs from the several delta sigma modulators and tomultiplex together the several delta sigma modulated outputs into apulse train.

In a further embodiment, the input signal is an orthogonalfrequency—division multiplexing (OFDM) modulated signal.

In a further embodiment again, the OFDM signal includes several symbolsand the signal demultiplexer demultiplexes the input signal using theseveral symbols and the signal multiplexer multiplexes the several deltasigma modulated outputs using the several symbols.

In another embodiment, the input signal is selected from the groupconsisting of a complex base-band signal, an RF signal, and a WiFibase-band signal.

In yet another embodiment, the delta sigma modulator includes aswitch-mode power amplifier configured to receive the pulse train forsignal amplification.

In still another embodiment, the delta sigma modulator includes afrequency up-converter configured to receive the pulse train.

In yet another embodiment again, a clock frequency of a delta sigmamodulator in the several delta sigma modulators is an integer divider ofthe pulse train output frequency

In a further embodiment, each delta sigma modulator in the several deltasigma modulators includes a noise shaping filter that is un-constrained.

In still a further embodiment again, each delta sigma modulator outputsa three level signal (−1, 0, 1) that drives a switch-mode poweramplifier (PA), where a ‘1’ means the switch-mode PA outputs a positivevoltage pulse, a ‘−1’ means the switch-mode PA outputs a negativevoltage pulse, and a ‘0’ means the PA is off.

In another embodiment again, an output of a delta sigma modulator in theseveral delta sigma modulators drives a linear amplifier, wherein theoutput is selected from the group consisting of a constant amplitudesignal and a zero state signal, wherein the zero state signal turns offamplitude.

In a still further embodiment, an output of a delta sigma modulator inthe several delta sigma modulators drives a switch-mode power amplifier(PA), wherein the output includes several discrete signal levels.

In another embodiment further, a delta sigma modulator design isun-constrained and the equivalent oversample ratio of the delta sigmamodulator is increased by N, where N equals the number of parallel deltasigma modulators.

In one embodiment, a switch-mode power amplifier system includes: asignal encoder including a delta sigma modulator; a switch-mode poweramplifier; a reconstruction filter; where the delta sigma modulatorincludes: a signal demultiplexer configured to receive an input signaland to demultiplex the input signal into a plurality of streams ofsymbols at symbol boundaries; a plurality of delta sigma modulators,where each delta sigma modulator is configured to receive a stream ofsymbols from the plurality of streams of symbols and to generate a deltasigma modulated output; and a signal multiplexer configured to receive aplurality of delta sigma modulated outputs from the plurality of deltasigma modulators and to multiplex together the plurality of delta sigmamodulated outputs into a pulse train.

In a further embodiment, the input signal is an orthogonalfrequency—division multiplexing (OFDM) modulated signal.

In a further embodiment still, the OFDM signal comprises a plurality ofsymbols; and the signal demultiplexer demultiplexes the input signalusing the plurality of symbols and the signal multiplexer multiplexesthe plurality of delta sigma modulated outputs using the plurality ofsymbols.

In still a further embodiment, the input signal is selected from thegroup consisting of a complex base-band signal, an RF signal, and a WiFibase-band signal.

In yet a further embodiment, the switch mode power amplifier systemincludes a frequency up-converter configured to receive the pulse train.

In another embodiment, a clock frequency of a delta sigma modulator inthe plurality of delta sigma modulators is an integer divider of thepulse train output frequency.

In another embodiment, each delta sigma modulator in the several deltasigma modulators includes a noise shaping filter that is un-constrained.

In still another embodiment, each delta sigma modulator outputs a threelevel signal (−1, 0, 1) that drives the switch-mode power amplifier(PA), wherein a ‘1’ means the switch-mode PA outputs a positive voltagepulse, a ‘−1’ means the switch-mode PA outputs a negative voltage pulse,and a ‘0’ means the PA is off.

In yet another further embodiment, an output of a delta sigma modulatorin the several delta sigma modulators drives the switch-mode poweramplifier (PA), wherein the output includes a plurality of discretesignal levels.

In still a further embodiment, a delta sigma modulator design isun-constrained and the equivalent oversample ratio of the delta sigmamodulator is increased by N, where N equals the number of parallel deltasigma modulators.

In one embodiment, a linear power amplifier system includes: a signalencoder comprising a delta sigma modulator; a linear power amplifier(PA); a reconstruction filter; where the delta sigma modulator includes:a signal demultiplexer configured to receive an input signal and todemultiplex the input signal into several streams of symbols at symbolboundaries; several delta sigma modulators, where each delta sigmamodulator is configured to receive a stream of symbols from the severalstreams of symbols and to generate a delta sigma modulated output; and asignal multiplexer configured to receive several delta sigma modulatedoutputs from the several delta sigma modulators and to multiplextogether the several delta sigma modulated outputs into a pulse train;and where an output of a delta sigma modulator in the several deltasigma modulators drives the linear power amplifier (PA), wherein theoutput includes a plurality of discrete signal levels.

In a further embodiment, the input signal is an orthogonalfrequency-division multiplexing (OFDM) modulated signal, where the OFDMsignal includes a plurality of symbols; and the signal demultiplexerdemultiplexes the input signal using the several symbols and the signalmultiplexer multiplexes the several delta sigma modulated outputs usingthe several symbols.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates an example architecture of an envelope tracking poweramplifier system.

FIG. 2 illustrates an example of a switch-mode PA system.

FIG. 3 illustrates an example of a delta sigma modulator implementedwith an error feedback architecture.

FIG. 4 illustrate an example of an OFDM modulator in accordance with anembodiment of the invention

FIG. 5 illustrates an example of an OFDM signal in a time domain inaccordance with an embodiment of the invention.

FIG. 6 illustrates a parallel delta sigma modulator architecture inaccordance with an embodiment of the invention.

FIG. 7 illustrates an example of a noise shaping filter designed for aWiFi system at carrier frequency 5.7 GHz in accordance with anembodiment of the invention.

FIG. 8 illustrates a WiFi switch-mode PA design in accordance with anembodiment of the invention.

FIG. 9 illustrates an example of a delta sigma modulator output.

FIG. 10 illustrates a histogram of three output levels in accordancewith an embodiment of the invention

FIG. 11 illustrates an output spectrum in accordance with an embodimentof the invention.

FIG. 12 illustrates a transmit signal constellation in accordance withan embodiment of the invention.

FIG. 13 illustrates a parallel delta sigma modulator with base-bandsignal input in accordance with an embodiment of the invention.

FIG. 14 illustrates a power amplifier architecture with frequencyup-converter in accordance with an embodiment of the invention.

FIG. 15 illustrates an output of a delta sigma modulator in accordancewith an embodiment of the invention.

FIG. 16 illustrates an application of a parallel delta sigma modulationarchitecture in an RF band in accordance with an embodiment of theinvention.

DETAILED DESCRIPTION

Turning now to the drawings, power amplifier (PA) systems and methodsfor achieving high efficiency in PA architectures for RF applications inaccordance with various embodiments of the invention are illustrated.

One of the most important parameters of an amplifier is its powerconversion efficiency. Power conversion efficiency is a measure of howeffectively an amplifier converts power drawn from a DC supply to usefulsignal (e.g., an RF signal) power delivered to a load. Power that is notconverted to useful signal power is typically dissipated as heat; andfor power amplifiers that have a low efficiency, the thermal andmechanical requirements resulting from high levels of heat dissipationare often limiting factors in their design.

As such, power amplifiers may be categorized according to the followingtwo groups as related to their power conversion efficiency (inadditional to the classes of amplifiers described above): 1) linearamplifiers, and 2) non-linear amplifiers. Linear amplifiers, as the nameimplies, maintain good signal linearity at the amplifier output.However, a linear amplifier typically has a lower power conversionefficiency compared to a non-linear amplifier. A non-linear amplifiercan achieve high power conversion efficiency at the expense of signallinearity degradation. It is traditionally used in communication systemsthat transmit signals with a constant amplitude envelope such as systemsthat employ frequency modulation (FM). In modern communication systemsthat use wider bandwidth and more bandwidth efficient modulation,linearity performance is important so linear amplifiers are almostalways used. On the other hand, bandwidth efficient modulationtranslates to higher peak to average power ratio (PAPR) for the signal.Linear PA power conversion efficiency typically degrades with theincrease of signal PAPR. For example, class A PA power conversionefficiency can be estimated as 10^((−PAPR/10)) assuming ideal circuitryfor the rest of the system. For a sinusoid signal that swings betweenthe voltage rails, the efficiency is typically 50% (PAPR is 3 dB). For atypical LTE or WiFi signal, the PAPR can be between 8 to 10 dB. At 10 dBPAPR, the power conversion efficiency is 10%. To output 250 mW of RFpower, the PA consumes 2.5 W where 2.25 W or 90% are dissipated as heat.

Several power amplifier architectures have been proposed to improvepower conversion efficiency. One example is an envelope tracking poweramplifier. Envelope tracking describes an approach to RF amplifierdesign in which the power supply voltage applied to the RF poweramplifier is continuously adjusted to increase the proportion of time atwhich the amplifier is operating at peak efficiency with respect to thepower required at each instant of transmission. An example architectureof an envelope tracking power amplifier system is illustrated in FIG. 1.One skilled in the art will appreciate that the power amplifier system100 may be embodied in hardware components, or a combination of hardwareand firmware and/or software components. Furthermore, one skilled in theart will appreciate that the power amplifier system 100 is only arepresentation of an envelope tracking power amplifier system and theexact components and/or processes performed may be different in variousother systems implementing envelope tracking.

As illustrated, the envelope tracking system 100 includes a main RF path101 and an envelope shaping signal generation path 102. The main RF path101 is where an in phase (I) signal 110 and quadrature (Q) signal 115can be used to create a composite RF signal 120 that is passed to the RFamplifiers 125.

Envelope shaping signal generation path 102 is a signal chain that cangenerate an envelope shaping signal. It consists of several components,including magnitude calculator 140, pre-envelope gain circuit 145,envelope shaping circuit 150, post-envelope gain/offset circuit 155,and/or DAC 160, to generate a signal appropriate to the operation of theamplifier and the prevailing signal conditions.

Envelope tracking modulator/supply 170 can modulate the voltage to thepower amplifier 125 so that the amplifier is operating at its maximumefficiency point.

With respect to delay balancing, the delays through the various signalpaths 101 and 102 can mean that the RF signal and the envelope shapingsignal each have their own delays. These delays may need to becompensated for to synchronize the RF envelope and the envelope trackingmodulator/supply.

While achieving good power conversion efficiency, there are severaldrawbacks for the envelope tracking PA architecture, including thefollowing:

-   -   Wide bandwidth requirement for the envelope tracking power        supply: The power supply tracking bandwidth typically needs to        be two to three times wider than the signal bandwidth. As the        communication system moves towards wider bandwidth channel for        higher throughput, this requirement becomes very difficult if        not impossible to achieve.    -   Mismatch between the main RF path and envelope signal path: This        mismatch can be time varying. To match these two paths at all        times, a background calibration algorithm with output monitoring        is typically required. This adds substantial complexity to the        overall PA system.

Recently, switch-mode PA architectures (e.g., classes D, E, F, and S,among others) have emerged to be a popular choice for high powerconversion efficiency PA design. A switch-mode PA may be compatible withdigital signal processing and ideally 100% efficient with a tuned outputload. An example of a switch-mode PA system is illustrated in FIG. 2.One skilled in the art will appreciate that the switch-mode PA system200 may be embodied in hardware components, or a combination of hardwareand firmware and/or software components. Furthermore, one skilled in theart will appreciate that the switch-mode PA system 200 is only arepresentation of a switch-mode PA system and the exact componentsand/or processes performed may be different in various other switch-modePA systems.

As illustrated in FIG. 2, switch-mode PA system 200 includes thefollowing three major components: 1) signal encoder which is typicallyimplemented using a band-pass delta sigma modulator 205; 2) switch-modepower amplifier 210; and 3) reconstruction filter or band-pass filter220. The signal encoder 205 converts the RF signal into a sequenceincluding a very fast switching pulse train. This pulse train can beused to control the switch-mode power amplifier 210. The switch-modepower amplifier 215 may be almost always operating in the saturationregion so it is very power efficient. The theoretical power conversionefficiency for an ideal switch-mode amplifier is 100%. Even though aswitch-mode power amplifier may be operating in a non-linear mode, itdoesn't affect the linearity of the output signal since the amplifierinput is a pulse train. The linearity of the output signal is determinedby the signal encoder.

Switch-mode PAs (class-D) have been very popular at audio frequencies,achieving close to its theoretically 100% conversion efficiency in realworld applications. Audio frequencies are generally lower than thefrequencies of RF transmissions and the typical Class-D PA switches at afew MHz. On the other hand, RF amplifiers in the microwave frequencyrange typically operate in the multi-GHz frequencies. As such, newclasses of switch-mode PAs (e.g., classes E, F, and S, among others)have been developed to achieve faster switching speed.

Achieving multi-GHz switching frequencies can be challenging for severalreasons. First, switch-mode power amplifiers generally need to switch ata frequency that is at least three times that of the output frequencyband and often switch at frequencies exceeding 10 GHz. With advances ofGallium nitrate (GaN) processes, switching speed of these magnitudeshave been made possible. Several companies, including Qorvo, Inc. andNorthrop Grumman Corporation offer GaN power transistor switches thatcan switch at frequencies exceeding multiple 10s of GHz at voltages upto 65V. Furthermore, a band-pass delta sigma modulator may need to runat a switching frequency at least two times that of the outputfrequency.

An error feedback architecture can be a popular implementation choicefor delta sigma modulation (e.g. for the purposes of performing digitalto analog conversion). As discussed further below, such delta sigmamodulators are not capable of switching at GHz frequencies. FIG. 3 is anexample of a delta sigma modulator implemented in accordance with anerror feedback architecture. As shown in FIG. 3, the quantized error canbe calculated every clock cycle and can be used to calculate the nextquantizer input. Thus, the speed of the delta sigma modulator is limitedby the delay of the feedback path calculation. This delay typicallydepends on the feedback transfer function complexity and the transistordelay. The transistor delay is not expected to be improved significantlygoing forward. Thus, the delta sigma modulator is limited to a fewhundred MHz. As noted above, however, use cases are envisaged inaccordance with various embodiments of the invention in whichswitch-mode PA delta sigma modulators are required to switch at speedsexceeding 10 GHz. One skilled in the art will recognize that only thecomponents of a delta sigma modulator 300 necessary for understandingthe error feedback path are shown in FIG. 3. Furthermore, the componentsand/or processes illustrated in FIG. 3 may be implemented by hardwarecomponents, or a combination of hardware and firmware and/or softwarecomponents.

As illustrated in FIG. 3, delta sigma modulator 300 includes an addercomponent 305 that receives a delta sigma modulator input signal 310 anda filtered error signal 350. The input signal 310 and the filtered errorsignal 350 are combined by the adder component 305 to output a correctedinput signal 330. The corrected input signal can be received by a 1-bittruncator component 315 and an adder component 325. The 1-bit truncatorcomponent 315 generates a delta sigma modulator output signal 320 thatis provided as an output and is received by adder component 325. Theadder component 325 combines the corrected input signal and the outputsignal to generate an error signal 340. H_(e) 335 is an error feedbacktransfer function. The quantizer error may be calculated and filtered bythe error feedback transfer function He 335 from the error signal togenerate the filtered error signal. The filtered error signal isprovided to adder component 305 to be summed with the delta sigmamodulator input signal 310 for quantization.

U.S. patent application Ser. No. 15/470,805 entitled “Systems andMethods for Fast Delta Sigma Modulation Using Parallel Path FeedbackLoops”, filed Mar. 27, 2017 discloses interpolated filter architecturesfor delta sigma modulator noise shaping filters, the disclosure of whichis herein incorporated by reference in its entirety. Using aninterpolated filter as the noise shaping filter may allow for a parallelimplementation for the delta sigma modulator at a lower clock speed.Most any arbitrary high-speed delta sigma modulator can be implementedusing the parallel architectures described in accordance with manyembodiments of the invention, as described in detail below. However,limiting the noise shaping filter choice to an interpolated filter canbe a major constraint for delta sigma modulator designs as it may bedifficult to design a noise shaping filter with both wide bandwidth andsharp rejection using the interpolated filter architectures.

Accordingly, many embodiments of the invention provide a parallel deltasigma modulator architecture implementation without an interpolatedfilter constraint on the noise shaping filter. Signal modulationstructures for parallel implementation are described in detail below.

Parallel Delta Sigma Modulator Architectures

Existing device technology typically limits switch-mode PA switchingspeeds to about 10 GHz. This limits the operation frequency of theswitch-mode PA to be a few GHz, whereas newer standards such as 5G arealready exploring millimeter wave frequency up to 100 GHz for morebandwidth. Accordingly, many embodiments of the delta sigma modulatorcan be extended to include a frequency up-converter in order to allowoperation in frequencies above 10 GHz. The frequency up-converted signalmay be amplified by a linear PA (class-A, A/B, B, and C). Given that thepost delta sigma modulator signal in accordance with many embodiments ofthe invention has a much lower PAPR or more favorable amplitudedistribution than the source signal, the PA efficiency can besignificantly improved.

In many embodiments, the parallel delta sigma modulation architecturecan be applied to most all PA classes (e.g., class A, A/B, B, C, D, E,F, S, among others) that amplify orthogonal frequency—divisionmultiplexing (OFDM) signals. Delta sigma modulation can be an effectiveway to reduce high PAPR resulting from OFDM modulation without degradingthe error vector magnitude (EVM) of the transmitted signal. Every dBreduction in PAPR can translate to a 1 dB improvement in PA efficiency.Accordingly, implementations of parallel delta sigma modulationarchitectures in accordance with certain embodiments of the inventioncan provide about a 3 dB reduction in PAPR or 100% improvement in PAefficiency.

Parallel Delta Sigma Modulator Architectures for OFDM

Most modern wideband communication systems are based on OFDM and itsvariants. For example, a 4G LTE downlink utilizes OFDMA and the uplinkis SC-FDMA. Likewise, most all high speed WiFi technologies, includingstandards 802.11a, 802.11n, 802.11ac among others, are based on OFDM.OFDM is also used in cable and terrestrial systems as well as manyemerging communication standards. FIG. 4 illustrate an example of anOFDM modulator in accordance with an embodiment of the invention. Bitsstream 405 can generate the payload bits and they can be mapped toconstellation points in symbol mapper 410. A serial/parallel 415 can beused to convert serial symbol stream to vector prior to IFFT 420. Aparallel/serial 425 can be used to convert IFFT output from vector formback to serial stream. Cyclic prefix can be added in add cyclic prefix430. A up-sample block 435 may be used to up-sample digital signal tohigher sample rate. The up-sample block output 436 can be in base-bandand can also be known as the base-band OFDM modulated signal. Thebase-band OFDM modulated signal can be up-converted to RF-band withmixer 440. The carrier can be generated by DDFS 445. The mixer output441 may be known as the RF band OFDM modulated signal and it can beconverted directly to RF signal with a high-speed DAC. Although FIG. 4illustrates a particular OFDM modulator, any of a variety of OFDMmodulators may be utilized as appropriate to the requirements ofspecific applications in accordance with embodiments of the invention.

FIG. 5 illustrates an example of an OFDM modulated signal in the timedomain. A key feature of OFDM-based systems is adding a cyclic prefix toa symbol. A cyclic prefix refers to the prefixing of a symbol with arepetition of the end. A cyclic prefix is added at the beginning of eachOFDM signal. FIG. 5 illustrates a cyclic prefix 505 added to a symbol N510, for each symbol N to symbol (N+L+1). The cyclic prefix is typicallydiscarded by a receiver. However, the cyclic prefix may serve severalpurposes.

First, the cyclic prefix can serve as a guard interval, whereby iteliminates the inter-symbol interference from a previous symbol. Also,as a repetition of the end of the symbol, it can allow the linearconvolution of a frequency-selective multipath channel to be modelled ascircular convolution, which in turn may be transformed to the frequencydomain using a discrete Fourier transform. This approach allows forsimple frequency-domain processing, such as channel estimation andequalization.

Power amplifiers in accordance with many embodiments of the inventionincorporate a parallel delta sigma modulator architecture for improvedpower conversion efficiency. An example of a parallel delta sigmamodulator architecture that can be employed in a power amplifier inaccordance with an embodiment of the invention is illustrated in FIG. 6.As illustrated in FIG. 6, the incoming OFDM modulated signal 605 isdemultiplexed using signal demultiplexer 610 into multiple streams 615of symbols 620. Each symbol 620 includes an up-sampled IFFT output andits cyclic prefix. Each signal stream 615 is processed by a delta sigmamodulator 630 running at a fraction of the full clock speed such thatthe delta-sigma modulator clock frequency can be an integer divider ofthe final pulse train output frequency. In some embodiments, the signaldemultiplexer may output symbols in parallel such that the output is atthe full clock speed and each delta-sigma modulator 630 (1 through L) inthe 1 through L parallel delta sigma modulators may use a clock signalthat is running at a clock speed of 1/L to read an input signal. Incertain embodiments, each delta sigma modulator 1 through L 630 may berunning at a staggered clock signal that is shifted by 1/L clock cyclesrelative to each other. For example, the signal demux 610 may output asymbol at a clock speed of 16 Ghz and each delta-sigma modulator may berunning a clock speed of 1 Ghz with a total of 16 delta-sigma modulatorsrunning in parallel. In some embodiments, the delta-sigma modulators 630may use the same clock signal. In certain embodiments, the delta-sigmamodulators 630 may each use a different clock signal that is staggeredrelative to each other. The amount that each clock signal may bestaggered relative to each other can be a phase shift of 1/L times thefull clock speed of the signal demux 610, where L is the number of deltasigma modulators running in parallel. For example, at a full clock speedof 16 GHz for the signal demux 610, each delta sigma modulator 1 throughL 630 can have a clock signal running at 1 Ghz, and each delta sigmamodulator can have a phase shift of 1/L clock signals relative to eachother. Thus, in this example, each delta sigma modulator would receive anew symbol at 1/16 the full clock speed of the signal demux 610.

The outputs of these delta sigma modulators 630 are multiplexed togetherusing signal multiplexer 640 into one pulse train 650. This pulse train650 can be fed into the switch-mode PA for signal amplification. In theparallel implementation illustrated in FIG. 6, the noise shaping filterof each delta sigma modulator can be un-constrained. This gives adesigner another degree of freedom to design a very wide bandwidth noiseshaping filter or a multi-band noise shaping filter. FIG. 7 illustratesan example of a noise shaping filter designed for a WiFi system atcarrier frequency 5.7 GHz.

The noise shaping filter

${H(z)} = {\frac{1 + {2.4754\mspace{11mu} z^{- 1}} + {3.5299\mspace{11mu} z^{- 2}} + {2.4754\mspace{11mu} z^{- 3}} + z^{- 4}}{1 + {2.2231\mspace{11mu} z^{- 1}} + {2.8504\mspace{11mu} z^{- 2}} + {1.8179\mspace{11mu} z^{- 3}} + {0.6660\mspace{11mu} z^{- 4}}}.}$

One skilled in the art will appreciate that the parallel delta sigmamodulator architecture illustrated in FIG. 6 may be embodied in embodiedin hardware components, or a combination of hardware and firmware and/orsoftware components. Furthermore, although FIG. 6 illustrates aparticular parallel delta sigma modulator architecture, any of a varietyof parallel delta sigma modulator architectures may be utilized asappropriate to the requirements of specific applications in accordancewith embodiments of the invention.

In many embodiments, the switch-mode PA design may be utilized with avariety of communication technologies, including WiFi standards 802.11a,802.11n, 802.11ac, among others. An example of a WiFi switch-mode PAdesign in accordance with an embodiment of the invention is illustratedin FIG. 8. As illustrated in FIG. 8, the WiFi base-band signal is firstup-converted using modulator and RF up-converter 805 to RF frequency inthe digital domain. A peak to average ratio (PAPR) component 810 can beused to control the peak signal power. The digital RF signal can beencoded with a delta sigma modulator 815. The delta sigma modulator 815outputs a three level signal (−1, 0, 1) that drives a switch-mode poweramplifier 820 to output an amplifier output 825. Although FIG. 8illustrates an example of a WiFi system, this design can be generalizedto any of a variety of OFDM based systems as appropriate to therequirements of specific applications in accordance with variousembodiments of the invention.

FIG. 9 illustrates an example of a delta sigma modulator output. Asillustrated in FIG. 9, ‘1’ means switch-mode PA outputs a positivevoltage pulse and ‘1’ means switch-mode PA outputs a negative voltagepulse. ‘0’ means the PA is off. In many embodiments, given the PA can beeither off or output a single level, no back off is needed and the powerconversion efficiency is very high.

FIG. 10 illustrates a histogram of the 3 output levels in accordancewith an embodiment of the invention. As shown in FIG. 10, the PA is offmost of the time and dissipates no power. The ‘0’ state (off state)reduces the DC power to the PA and improves the overall powerefficiency. In many embodiments, the power encoder efficiency can bedefined as the ratio of the desired signal power and the total power. Inthe case illustrated in FIG. 10, the power encoder efficiency after thedelta sigma modulator is 56%. That is 3 times the input signal. Giventhe bulk of error power is out of band, a tuned PA design can be used tosuppress the out-of-band energy. In theory, the power conversionefficiency of this tuned switch-mode PA can achieve 100%. The outputspectrum is illustrated in FIG. 11 and the transmit signal constellationis illustrated in FIG. 12.

In many embodiments, the parallel delta sigma modulation architecturecan also be applied at a base-band signal. Performing delta sigmamodulation in the base-band may allow for higher frequency conversionsuch that the final modulated signal can be significantly higher (e.g.,higher than 10 GHz). A parallel delta sigma modulator with base-bandsignal input in accordance with an embodiment of the invention isillustrated in FIG. 13. The incoming base-band OFDM modulated signal1305 is demultiplexed using a signal demultiplexer 1310 into multiplestreams 1311 of symbols 1305. Each signal stream 1311 is processed by adelta sigma modulator 1315 running at a fraction of the full clockspeed. The outputs of these delta sigma modulators 1315 can bemultiplexed together using a signal multiplexer 1320 into one pulsetrain 1330. This pulse train 1330 can be feed into the frequencyupconverter for RF conversion. As illustrated in FIG. 13, the deltasigma modulator 1315 full clock speed should be an integer divider ofthe final carrier frequency. For example, if the transmitter carrierfrequency is 5.7 GHz, possible choices for the delta sigma modulator1315 full clock speed include 5.7 GHz, 2.85 GHz, 1.9 GHz, or otherinteger divisions of 5.7 GHz. The delta sigma modulator clock speed ischosen depends on performance requirement of the system. Although FIG.13 illustrates a parallel delta sigma modulation architecture applied ata base-band signal, the parallel delta sigma modulation architecture maybe applied to a variety of different input signals as appropriate to therequirements of specific applications in accordance with variousembodiments of the invention.

In many embodiments, the PA architecture may use a frequencyup-converter to increase power conversion efficiency. An example of a PAarchitecture that utilizes a frequency up-converter in accordance withan embodiment of the invention is illustrated in FIG. 14. The base-bandsignal 1401 is processed by the PAPR reduction component 1405. It isthen encoded with a delta sigma modulator 1410. The delta sigmamodulator 1410 can output either 0 or a constant power signal. In thisexample, 6 equal spaced constellation points on a circle are chosen.Although FIG. 14 illustrates a particular PA architecture with afrequency up-converter, any of a variety of PA architectures may beutilized as appropriate to the requirements of specific applications inaccordance with various embodiments of the invention.

An example of an output of the delta sigma modulator illustrated in FIG.14 in accordance with an embodiment is illustrated in FIG. 15. Asillustrated, ‘0’ means the PA is off or no signal is transmitted.Non-zero means the PA is on. Given the PA is either off or has as itsoutput a constant amplitude signal, the power conversion efficiency isvery high. The ‘0’ state (off state) reduces the DC power to the PA andimproves the overall power efficiency. Given the bulk of error power isout of band, a tuned PA design can be used to suppress the out-of-bandenergy. Using a class-B PA as an example, the power conversionefficiency of amplifying this signal can achieve the theoretical limitof class-B or 78%.

In many embodiments, the parallel delta sigma modulation architecturecan be applied to a variety of different applications. An example of anapplication of the parallel delta sigma modulation architecture in theRF band in accordance with an embodiment of the invention is illustratedin FIG. 16. As illustrated in FIG. 16, the delta sigma modulator 1610 isworking on the signal in the RF band with the PAPR reduction component1610. A variety of processes can be utilized for performing PAPRreduction and most all of them generate noise. In many embodiments,delta sigma modulation can suppress the in-band noise at the expense ofout-of-band noise growth. In many embodiments, by having a delta sigmamodulator work together with a PAPR reduction component results in lowerPAPR without SNR degradation of the desired signal. When amplifying thissignal with a linear PA such as a class A PA, 1 dB reduction in PAPR cantranslate to a 1 dB gain in power conversion efficiency. Although FIG.16 illustrates a particular parallel delta sigma modulation architecturein the RF band, any of a variety of parallel delta sigma modulationarchitectures in any of a variety of signal bands may be utilized asappropriate to the requirements of specific applications in accordancewith various embodiments of the invention.

The various high efficiency power amplifier architectures describedabove have many advantages over existing PA designs. In particular, theparallel delta sigma modulation architectures of many embodiments makespossible very high-speed delta sigma modulation on the RF signal itself.This enables switch-mode PA designs for multi-GHz band. Furthermore, inmany embodiments there is a ‘0’ state in the multi-level delta sigmaoutput which turns off the PA to save power. Combined with tuned outputloading, a switch-mode PA in accordance with many embodiments of theinvention can approach 100% efficiency.

Likewise, a switch-mode PA in accordance with many embodiments of theinvention is feed-forward and thus there is no feedback monitoringneeded as in, for example, an envelope tracking PA architecture.Accordingly, the design complexity of many embodiments of theswitch-mode PA can be reduced significantly.

In addition, in many embodiments, the parallel delta sigma modulationarchitecture can be paired with a frequency up-converter to generatehigher frequency output. The delta sigma modulator in accordance withmany embodiments of the invention can output either ‘0’ or a constantamplitude signal. Having a constant amplitude signal may allow a linearPA to operate at its peak efficiency. The ‘0’ state may allow the PA toshut off and save power. In many embodiments of the invention, using adelta sigma architecture with a tuned class-B PA, the power conversionefficiency can approach the peak efficiency of a class-B amplifier or78%.

In many embodiments, the parallel delta sigma modulation architecturecan be paired with a PAPR reduction block to get lower PAPR ratio. Inmany embodiments, the delta sigma modulator can suppress in-band noiseat the expense of out-of-band noise growth. This may allow higher noisetolerance from the PAPR reduction and reduces the final PAPR ratio. Thispower conversion efficiency gain from PAPR reduction can be applied toPAs of any class appropriate to the requirements of a given application.

In many embodiments, the parallel delta sigma modulation architectureallows parallel computation of the feedback path and thus is able toachieve faster effective speed with parallelization. In severalembodiments, the parallel delta sigma modulation architecture also takesadvantage of the guard interval between OFDM symbols in the OFDMmodulation. Multiple delta sigma processors can be used to processmultiple OFDM symbols at the same time. Furthermore, in manyembodiments, the delta sigma modulator may no longer be limited by thedevice delay of the feedback path. Accordingly, the number ofcomputations can be the same as serial computation with no overhead forparallelization.

In many embodiments, the parallel delta sigma modulation architecturecan achieve very high-speed delta sigma modulator speed with applicationin direct-RF conversion, switched-mode PA, and PAPR reduction, amongvarious other applications. Although the present invention has beendescribed in certain specific aspects, many additional modifications andvariations would be apparent to those skilled in the art. It istherefore to be understood that the present invention may be practicedotherwise than specifically described, including various changes in theimplementation. Thus, embodiments of the present invention should beconsidered in all respects as illustrative and not restrictive.

What is claimed is:
 1. A parallel delta sigma modulator comprising: asignal demultiplexer configured to receive an input signal and todemultiplex the input signal into a plurality of streams of symbols atsymbol boundaries; a plurality of delta sigma modulators, where eachdelta sigma modulator is configured to receive a stream of symbols fromthe plurality of streams of symbols and to generate a delta sigmamodulated output; and a signal multiplexer configured to receive aplurality of delta sigma modulated outputs from the plurality of deltasigma modulators and to multiplex together the plurality of delta sigmamodulated outputs into a pulse train.
 2. The parallel delta sigmamodulator of claim 1, wherein the input signal is an orthogonalfrequency—division multiplexing (OFDM) modulated signal.
 3. The paralleldelta sigma modulator of claim 2, wherein: the OFDM signal comprises aplurality of symbols; and the signal demultiplexer demultiplexes theinput signal using the plurality of symbols and the signal multiplexermultiplexes the plurality of delta sigma modulated outputs using theplurality of symbols.
 4. The parallel delta sigma modulator of claim 1,wherein the input signal is selected from the group consisting of acomplex base-band signal, an RF signal, and a WiFi base-band signal. 5.The parallel delta sigma modulator of claim 1, further comprising aswitch-mode power amplifier configured to receive the pulse train forsignal amplification.
 6. The parallel delta sigma modulator of claim 1,further comprising a frequency up-converter configured to receive thepulse train.
 7. The parallel delta sigma modulator of claim 1, wherein aclock frequency of a delta sigma modulator in the plurality of deltasigma modulators is an integer divider of the pulse train outputfrequency
 8. The parallel delta sigma modulator of claim 1, wherein eachdelta sigma modulator in the plurality of delta sigma modulatorscomprises a noise shaping filter that is un-constrained.
 9. The deltasigma modulator of claim 1, wherein each delta sigma modulator outputs athree level signal (−1, 0, 1) that drives a switch-mode power amplifier(PA), wherein a ‘1’ means the switch-mode PA outputs a positive voltagepulse, a ‘−1’ means the switch-mode PA outputs a negative voltage pulse,and a ‘0’ means the PA is off.
 10. The delta sigma modulator of claim 1,wherein an output of a delta sigma modulator in the plurality of deltasigma modulators drives a linear amplifier, wherein the output isselected from the group consisting of a constant amplitude signal and azero state signal, wherein the zero state signal turns off amplitude.11. The delta sigma modulator of claim 1, wherein an output of a deltasigma modulator in the plurality of delta sigma modulators drives aswitch-mode power amplifier (PA), wherein the output comprises aplurality of discrete signal levels.
 12. The delta sigma modulator ofclaim 1, wherein a delta sigma modulator design is un-constrained andthe equivalent oversample ratio of the delta sigma modulator isincreased by N, where N equals the number of parallel delta sigmamodulators.
 13. A switch-mode power amplifier system comprising: asignal encoder comprising a delta sigma modulator; a switch-mode poweramplifier; a reconstruction filter; wherein the delta sigma modulatorcomprises: a signal demultiplexer configured to receive an input signaland to demultiplex the input signal into a plurality of streams ofsymbols at symbol boundaries; a plurality of delta sigma modulators,where each delta sigma modulator is configured to receive a stream ofsymbols from the plurality of streams of symbols and to generate a deltasigma modulated output; and a signal multiplexer configured to receive aplurality of delta sigma modulated outputs from the plurality of deltasigma modulators and to multiplex together the plurality of delta sigmamodulated outputs into a pulse train.
 14. The switch-mode poweramplifier of claim 13, wherein the input signal is an orthogonalfrequency—division multiplexing (OFDM) modulated signal.
 15. Theswitch-mode power amplifier of claim 14, wherein: the OFDM signalcomprises a plurality of symbols; and the signal demultiplexerdemultiplexes the input signal using the plurality of symbols and thesignal multiplexer multiplexes the plurality of delta sigma modulatedoutputs using the plurality of symbols.
 16. The switch-mode poweramplifier of claim 13, wherein the input signal is selected from thegroup consisting of a complex base-band signal, an RF signal, and a WiFibase-band signal.
 17. The switch-mode power amplifier of claim 13,further comprising a frequency up-converter configured to receive thepulse train.
 18. The switch-mode power amplifier of claim 13, wherein aclock frequency of a delta sigma modulator in the plurality of deltasigma modulators is an integer divider of the pulse train outputfrequency
 19. The switch-mode power amplifier of claim 13, wherein eachdelta sigma modulator in the plurality of delta sigma modulatorscomprises a noise shaping filter that is un-constrained.
 20. Theswitch-mode power amplifier of claim 13, wherein each delta sigmamodulator outputs a three level signal (−1, 0, 1) that drives theswitch-mode power amplifier (PA), wherein a ‘1’ means the switch-mode PAoutputs a positive voltage pulse, a ‘−1’ means the switch-mode PAoutputs a negative voltage pulse, and a ‘0’ means the PA is off.
 21. Theswitch-mode power amplifier of claim 13, wherein an output of a deltasigma modulator in the plurality of delta sigma modulators drives theswitch-mode power amplifier (PA), wherein the output comprises aplurality of discrete signal levels.
 22. The switch-mode power amplifierof claim 13, wherein a delta sigma modulator design is un-constrainedand the equivalent oversample ratio of the delta sigma modulator isincreased by N, where N equals the number of parallel delta sigmamodulators.
 23. A linear power amplifier system comprising: a signalencoder comprising a delta sigma modulator; a linear power amplifier(PA); a reconstruction filter; wherein the delta sigma modulatorcomprises: a signal demultiplexer configured to receive an input signaland to demultiplex the input signal into a plurality of streams ofsymbols at symbol boundaries; a plurality of delta sigma modulators,where each delta sigma modulator is configured to receive a stream ofsymbols from the plurality of streams of symbols and to generate a deltasigma modulated output; and a signal multiplexer configured to receive aplurality of delta sigma modulated outputs from the plurality of deltasigma modulators and to multiplex together the plurality of delta sigmamodulated outputs into a pulse train; and wherein an output of a deltasigma modulator in the plurality of delta sigma modulators drives thelinear power amplifier (PA), wherein the output comprises a plurality ofdiscrete signal levels.
 24. The linear power amplifier of claim 23,wherein the input signal is an orthogonal frequency—divisionmultiplexing (OFDM) modulated signal, wherein: the OFDM signal comprisesa plurality of symbols; and the signal demultiplexer demultiplexes theinput signal using the plurality of symbols and the signal multiplexermultiplexes the plurality of delta sigma modulated outputs using theplurality of symbols.